Usually, in such capacitive sensors for measuring a physical parameter, the mobile common electrode forms part of an armature resiliently held between the two fixed electrodes. This common electrode is capable of moving a certain distance in the direction of one or the other of the fixed electrodes, via the action of a force, for example. In the inoperative state, the common electrode is ideally equidistant from both fixed electrodes, which defines equal capacitive values for the two capacitors. When the common electrode moves via the action, for example, of a force, the capacitive value of each capacitor varies inversely. An interface connected to the electronic circuit capacitive sensor is for providing an output signal in the form of a voltage that depends upon the variation in the capacitances of the capacitors.
In an ideal case, the output voltage varies in a linear manner in relation to the movement of the mobile common electrode. However, since the electronic circuit parts are made in the form of at least one integrated circuit in a semiconductor substrate, stray capacitances, which are added to the capacitor capacitances, must be taken into account. These stray capacitances are virtually independent of the movement of the common electrode, which creates non-linearities. Consequently, the electronic circuit output voltage does not vary linearly in relation to the movement of the mobile common electrode. These stray capacitors also have the effect of lowering the sensitivity or yield of the electronic circuit.
Since the MEMS type sensor can also be integrated in a semiconductor substrate, such as a silicon substrate, there is also a problem of non-linearity also linked to the potential of the substrate during operation of the sensor. The substrate potential is difficult to control across the entire structure of said sensor, since the substrate is never totally conductive. Because of this non-linearity, the measured electrostatic force is not zero when the electronic circuit is in inoperative mode. The influence of the substrate potential on the electrostatic force leads to a variation in the measured real force, which is applied across the moving common electrode, which is a drawback of the electronic circuit. Moreover, stray capacitors in parallel with the sensor's capacitors must also be taken into account. The capacitances of the stray capacitors are assumed to be quasi-constant and independent of the force applied to the electronic circuit. This has the effect of decreasing the sensitivity or yield of the electronic circuit, which is another drawback of the electronic circuit with a capacitive sensor.
In order, generally, to take a force, acceleration or pressure measurement, the fixed electrodes of the two capacitors are biased or excited cyclically by voltages of opposite polarity relative to an inoperative reference voltage. By biasing or polarising the two fixed electrodes at different voltage levels, the charge difference across the moving electrode can be measured and converted into an electronic circuit output voltage. When the output voltage has stabilised at its final value, the total charge across the moving electrode becomes zero. This output voltage can be supplied, sampled, to a processing circuit able to provide acceleration, force, pressure or angular velocity data, depending upon the structure of the sensor.
An electronic circuit with a capacitive sensor of the prior art is shown in FIG. 1, and the activation thereof is illustrated by a time diagram of various voltage signals in FIG. 2. The electronic circuit shown is based on an electronic circuit described in the article by Messrs. H Leuthold and F.
Rudolph, which appeared in the journal entitled, “Sensors and actuators” A21-23 (1990), pages 278 to 281.
The electronic circuit 1 shown includes an interface connected to a capacitive sensor 2, which includes two capacitors mounted in differential C1 and C2. The two capacitors have a common electrode Cm that can move between two fixed electrodes. The interface of electronic circuit 1 includes a charge transfer amplifier unit 4, which is connected at input to common electrode Cm, an integrator unit 5 for permanently supplying at output a voltage Vm equal to the integral of charges supplied by amplifier unit 4, and an excitation unit 3 for cyclically biasing or polarizing the fixed electrodes at determined voltage levels.
Excitation unit 3 includes four switches 12, 13, 14 and 15, which can be formed by MOS switching transistors in the integrated circuit. The first switch 12 is arranged between the output of integrator unit 5 and the fixed electrode of capacitor C1. The second switch 13 is arranged between the integrator unit output and the fixed electrode of capacitor C2. The third switch 14 is arranged between the high voltage terminal VDD of a continuous voltage source and the fixed electrode of capacitor C1. Finally, the fourth switch 15 is arranged between the low voltage terminal VSS of the voltage source and the fixed electrode of capacitor C2.
In the electronic circuit operating mode, each successive measuring period or cycle is divided into two phases P1 and P2 as shown in FIG. 2. The change from one phase to another is controlled by clock signals that are not shown, for respectively opening or closing the switches. Switches 12 and 13 are closed by signals SW2 at the “1” state in the first phase designated P1 in FIGS. 1 and 2, whereas switches 14 and 15 are open in this first phase P1. In this first phase P1, voltage Vm present at the integrator unit output is applied to each electrode of the sensor to discharge the two capacitors completely as shown by voltage diagrams VC1, VCm and VC2.
Switches 14 and 15 are closed by signals SW1 at the “1” state in the second phase designated P2, whereas switches 12 and 13 are open. In this second phase P2, voltage VDD is applied to the fixed electrode C1 seen in the VC1 diagram, whereas voltage VSS is applied to the fixed electrode C2 seen in the VC2 diagram. If the moving electrode is moved a certain distance in the direction of one or other of the fixed electrodes, the capacitances of the capacitors will vary inversely. This will lead to a difference in the charges accumulated by each capacitor, which also depends upon the voltage Vm previously applied to each electrode of the capacitors.
The final value of voltage Vm at the integrator unit output is obtained after several operating cycles of the electronic circuit as a function of the movement of the mobile electrode between the two fixed electrodes as shown in the VCm voltage diagram. In this case, the common electrode is moved in the direction of the fixed electrode of capacitor C1, which results in a final integrator unit output voltage, which is above the medium or intermediate voltage (VDD−VSS)/2. The potential of the common electrode has thus been adjusted to cancel out any charge flow and thus to maintain the total charge at zero in accordance with the principle of charge compensation.
For the operation of transferring charges accumulated by common electrode Cm, the charge transfer amplifier unit 4 includes an operational amplifier 10, three capacitors C3, C4 and C5 and two switches 16 and 17. The inverter input of this amplifier is connected to common electrode Cm. Capacitor C3 in parallel with switch 16 is connected between the inverter input and the output of amplifier 10. Capacitor C4 is connected between the output of amplifier 10 and the input of integrator unit 5. Capacitor C5 is connected between the non-inverter input and a reference voltage terminal Vref, which can be defined as earth DC equal to VSS or (VDD−VSS)/2 or to another potential. Finally, switch 17 is arranged between the output of integrator unit 5 and the non-inverter input of amplifier 10.
In the electronic circuit operating mode, the two switches 16 and 17 are closed by signals SW2 at the “1” state in first phase P1 to partly discharge capacitor C3 and polarise capacitor C5 with output voltage Vm at the non-inverter input of the amplifier. Voltage level Vm of capacitor C5 is maintained during second phase P2.
Integrator unit 5, which follows the charge transfer amplifier unit 4, includes two input switches 18 and 19, an operational amplifier 11 and an integration capacitor Cf. This capacitor Cf is connected between the inverter input and the output of amplifier 11, which supplies output voltage Vm of integrator 5. Input switch 18 is arranged between the output terminal of capacitor C4 of charge transfer unit 4 and the non-inverter input of amplifier 11. The potential of this non-inverter input of amplifier 11 is set at reference voltage Vref. Switch 19 is arranged between the output terminal of capacitor C4 of charge transfer unit 4 and the inverter input of amplifier 11.
In the electronic circuit operating mode, switch 18 is closed by signals SW2 at the “1” state in first phase P1 so that the voltage at the terminals of capacitor C4 of the charge transfer unit is equal to Vm if reference voltage Vref is at earth. Switch 19 is closed by signals SW1 at the “1” state in second phase P2 to perform a charge flow between the output terminal of capacitor C4 of charge transfer unit 4 and integrator 5. This charge flow from charge transfer amplifier unit 4 is integrated in capacitor Cf. Thus, the output voltage Vm is updated, i.e. altered by a quantity proportional to the charge accumulated across the common moving electrode during the second phase.
The operation of the electronic circuit described above is asymmetrical, since the fixed electrode of capacitor C1 is always polarized at the same potential VDD in each second phase P2, whereas the fixed electrode of capacitor C2 is always biased at VSS in each second phase P2. This type of integrated electronic circuit thus encounters the same problems of non-linearity mentioned above with reference to stray capacitors and the substrate potential, which is a drawback. Moreover, since the electronic circuit is made in the form of an integrated circuit, any voltage offset linked to unmatched electronic components cannot be removed, which is another drawback.
One way of improving non-linearities in the electronic circuit with a capacitive sensor was proposed in FR Patent No. 2 720 510, on which the electronic circuit of this invention is based. The difference between the electronic circuit presented here and that described above with reference to FIG. 1, is that it advantageously includes another compensation capacitor Cc placed at the integrator unit input. This compensation capacitor mainly compensates for the effects of stray capacitors, particularly those of the capacitive sensor, to increase the gain of the electronic circuit. However, this electronic circuit cannot prevent the electronic circuit output from being blocked or locked at the high potential or low potential of the supply voltage source after an abrupt variation, such as a shock applied to the sensor, outside the electronic circuit measuring range. This causes saturation of the electronic circuit interface, which is a drawback. Even if the high amplitude disappears, the interface saturation remains permanent, which means that the electronic circuit is no longer functional. It is therefore necessary to initialise the electronic circuit completely, for it to operate properly, which involves a relatively large set up time and greater electric power consumption.